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1、CHAPTER2: FREQUENCY CONVERTERS81 Control circuit The control circuit, or control card, is the fourth main compo- nent of the frequency converter and has four essential tasks: control of the frequency converter semi-conductors. data exchange between the frequency converter and periph- erals. gatherin
2、g and reporting fault messages. carrying out of protective functions for the frequency convert- er and motor. Micro-processors have increased the speed of the control circuit, significantly increasing the number of applications suitable for drives and reducing the number of necessary calculations. W
3、ith microprocessors the processor is integrated into the fre- quency converter and is always able to determine the optimum pulse pattern for each operating state. Fig. 2.29 shows a PAM-controlled frequency converter with intermediate circuit chopper. The control circuit controls the chopper (2) and
4、the inverter (3). U f 123 Fig. 2.29The principle of a control circuit used for a chopper- controlled intermediate circuit Control circuit for chopper frequency PI voltage regulator Sequence generator Control circuit for PAM frequency converter This is done in accordance with the momentary value of t
5、he intermediate circuit voltage. The intermediate circuit voltage controls a circuit that functions as an address counter in the data storage. The storage has the output sequences for the pulse pattern of the inverter. When the intermediate circuit voltage increases, the counting goes faster, the se
6、quence is completed faster and the output frequency increases. With respect to the chopper control, the intermediate circuit voltage is first compared with the rated value of the reference signal a voltage signal. This voltage signal is expected to give a correct output voltage and frequency. If the
7、 reference and intermediate circuit signals vary, a PI-regulator informs a cir- cuit that the cycle time must be changed. This leads to an adjustment of the intermediate circuit voltage to the reference signal. PAM is the traditional technology for frequency inverter control. PWM is the more modern
8、technique and the following pages detail how Danfoss has adapted PWM to provide particular and specific benefits. Danfoss control principle Fig. 2.30 gives the control procedure for Danfoss inverters. The control algorithm is used to calculate the inverter PWM switching and takes the form of a Volta
9、ge Vector Control (VVC) for voltage-source frequency converters. 82CHAPTER2: FREQUENCY CONVERTERS Fig. 2.30Control principles used by Danfoss SoftwareHardware (ASIC)Inverter VVCSynchronous 60 PWMMotor VVCplusAsynchronous SFAVM 60 PWM Control algorithmPWM VVC controls the amplitude and frequency of t
10、he voltage vector using load and slip compensation. The angle of the voltage vec- tor is determined in relation to the preset motor frequency (ref- erence) as well as the switching frequency. This provides: full rated motor voltage at rated motor frequency (so there is no need for power reduction) s
11、peed regulation range: 1:25 without feedback speed accuracy: 1% of rated speed without feedback robust against load changes A recent development of VVC is VVCplusunder which. The ampli- tude and angle of the voltage vector, as well as the frequency, is directly controlled. In addition to the propert
12、ies of VVC , VVCplusprovides: improved dynamic properties in the low speed range (0 Hz-10 Hz). improved motor magnetisation speed control range: 1:100 without feedback speed accuracy: 0.5% of the rated speed without feedback active resonance dampening torque control (open loop) operation at the curr
13、ent limit CHAPTER2: FREQUENCY CONVERTERS83 VVC control principle Under VVC the control circuit applies a mathematical model, which calculates the optimum motor magnetisation at varying motor loads using compensation parameters. In addition the synchronous 60 PWM procedure, which is inte- grated into
14、 an ASIC circuit, determines the optimum switching times for the semi-conductors (IGBTs) of the inverter. The switching times are determined when: The numerically largest phase is kept at its positive or nega- tive potential for 1/6of the period time (60). The two other phases are varied proportiona
15、lly so that the resulting output voltage (phase-phase) is again sinusoidal and reaches the desired amplitude (Fig. 2.32). 84CHAPTER2: FREQUENCY CONVERTERS 0,00 0,5 UDC 0,5 UDC 360060 60 120180240300 Fig. 2.31Synchronous 60 PWM (Danfoss VVC control) of one phase UDC= intermediate circuit voltage CHAP
16、TER2: FREQUENCY CONVERTERS85 Unlike sine-controlled PWM, VVC is based on a digital genera- tion of the required output voltage. This ensures that the fre- quency converter output reaches the rated value of the supply voltage, the motor current becomes sinusoidal and the motor operation corresponds t
17、o those obtained in direct mains connec- tion. Optimum motor magnetisation is obtained because the fre- quency converter takes the motor constants (stator resistance and inductance) into account when calculating the optimum output voltage. As the frequency converter continues to measure the load cur
18、- rent, it can regulate the output voltage to match the load, so the motor voltage is adapted to the motor type and follows load con- ditions. 0,00 0,50 1,00 0,50 1,00 U-VV-WW-U 360060120180240300 Fig. 2.32With the synchronous 60 PWM principle the full output voltage is obtained directly Switching p
19、attern of phase U Phase voltage (0-point half the intermediate circuit voltage) Combined voltage to motor VVCpluscontrol principle The VVCpluscontrol principle uses a vector modulation principle for constant, voltage-sourced PWM inverters. It is based on an improved motor model which makes for bette
20、r load and slip compensation, because both the active and the reactive current components are available to the control system and controlling the voltage vector angle significantly improves dynamic perfor- mance in the 0-10 Hz range where standard PWM U/F drives typically have problems. The inverter
21、 switching pattern is calculated using either the SFAVM or 60 AVM principle, to keep the pulsating torque in the air gap very small (compared to frequency converters using synchronous PWM). The user can select his preferred operating principle, or allow the inverter to choose automatically on the ba
22、sis of the heat- sink temperature. If the temperature is below 75C, the SFAVM principle is used for control, while above 75 the 60 AVM prin- ciple is applied. Table 2.01 gives a brief overview of the two principles: The control principle is explained using the equivalent circuit diagram (Fig. 2.33)
23、and the basic control diagram (Fig. 2.34). It is important to remember that in the no-load state, no current flows in the rotor (i= 0), which means that the no-load voltage can be expressed as: U = U L= (RS+ jSLS) is 86CHAPTER2: FREQUENCY CONVERTERS Table 2.01Overview: SFAVM versus 60 AVM Max. switc
24、hing Selectionfrequency ofProperties inverter SFAVMMax. 8 kHz1. low torque ripple compared to the synchronous 60 PWM (VVC) 2. no “gearshift” 3. high switching losses in inverter 60-AVMMax. 14 kHz1. reduced switching losses in inverter (by 1/3 compared to SFAVM) 2. low torque ripple compared to the s
25、ynchronous 60 PWM (VVC) 3. relatively high torque ripple compared to SFAVM in which: RSis the stator resistance, isis the motor magnetisation current, LSis the stator leakage inductance, Lhis the main inductance, LS(=LS+ Lh) is the stator inductance, and s(=2fs) is the angular speed of the rotating
26、field in the air gap The no-load voltage (UL) is determined by using the motor data (rated voltage, current, frequency, speed). Under a load, the active current (iw) flows in the rotor. In order to enable this current, an additional voltage (UComp) is placed at the disposal of the motor: The additio
27、nal voltage UCompis determined using the no-load and active currents as well as the speed range (low or high speed). The voltage value and the speed range are then deter- mined on the basis of the motor data. CHAPTER2: FREQUENCY CONVERTERS87 iw LR Rr Lh is LSRS + UL Uq UComp Fig. 2.33bEquivalent cir
28、cuit diagram for three-phase AC motors (loaded) iw LR Rr is LS Lh RS U = UL Uq Fig. 2.33aEquivalent circuit diagram of three-phase AC motor loaded) 88CHAPTER2: FREQUENCY CONVERTERS ffrequency (internal) fspreset reference frequency fcalculated slip frequency ISXreactive current components (calculate
29、d) ISYactive current components (calculated) ISX0, ISY0no- load current of x and y axes (calculated) Iu, Iv, Iwcurrent of phases U, V and W (measured) Rsstator resistance Rrrotor resistance angle of the voltage vectors 0no- load value theta load-dependent part of theta (compensation) TCTemperature o
30、f heat conductor/ heat sink UDCvoltage of DC intermediate circuit ULno- load voltage vector USstator voltage vector UCompload- dependent voltage compensation Umotor supply voltage Xhreactance X1stator leakage reactance X2rotor leakage reactance sstator frequency LSstator inductance LSsstator leakage
31、 inductance LRsrotor leakage inductance ismotor phase current (apparent current) iwactive (rotor) current Explanations for Fig. 2.33 (page 87) and Fig. 2.34 (page 89) xy PWM-ASIC ab xy ab 2 3 ISX0ISY0 ISX ISY Iu Iv Iw UL UDCTC I0 0 L fs f f p UU Ucomp f f f = 3 Rectifier Interver Motor Switching log
32、ic Voltage vector generator (no load) Load compen- sator Slip compen- sation Ramp Mains Basis VVCplus Motor- model CHAPTER2: FREQUENCY CONVERTERS89 Fig. 2.34Basis of VVCpluscontrol 90CHAPTER2: FREQUENCY CONVERTERS As shown in Fig. 2.34, the motor model calculates the rated no- load values (currents
33、and angles) for the load compensator (ISX0, Isyo) and the voltage vector generator (Io, o). Knowing the actual no load values makes it possible to estimate the motor shaft torque load much more accurately. The voltage vector generator calculates the no-load voltage vec- tor (UL) and the angle (L) of
34、 the voltage vector on the basis of the stator frequency, no-load current, stator resistance and inductance (see Fig. 2.33a). The resulting voltage vector ampli- tude is a composite value having added start voltage and load compensation voltage. The voltage vector Lis the sum of four terms, and is a
35、n absolute value defining the angular position of the voltage vector. As the resolution of the theta components () and the stator fre- quency (F) determines the output frequency resolution, the val- ues are represented in 32 bit resolution. One () theta compo- nent is the no load angle which is incl
36、uded in order to improve the voltage vector angle control during acceleration at low speed. This results in a good control of the current vector since the torque current will only have a magnitude which corre- sponds to the actual load. Without the no load angle component the current vector would te
37、nd to increase and over magnetise the motor without producing torque. The measured motor currents (Iu, Ivand Iw) are used to calcu- late the reactive current (ISX) and active current (ISY) compo- nents. Based on the calculated actual currents and the values of the voltage vector, the load compensato
38、r estimates the air gap torque and calculates how much extra voltage (UComp) is required to maintain the magnetic field level at the rated value. The angle deviation () to be expected because of the load on the motor shaft is corrected. The output voltage vector is repre- sented in polar form (p). T
39、his enables a direct overmodulation and facilitates the linkage to the PWM-ASIC. The voltage vector control is very beneficial for low speeds, where the dynamic performance of the drive can be significant- ly improved, compared to V/f control by appropriate control of the voltage vector angle. In ad
40、dition, steady stator performance is obtained, since the control system can make better estimates for the load torque, given the vector values for both voltage and current, than is the case on the basis of the scalar signals (amplitude values). CHAPTER2: FREQUENCY CONVERTERS91 Field-oriented (Vector
41、) control Vector control can be designed in a number of ways. The major difference is the criteria by which the active current, magne- tising current (flux) and torque values are calculated. Comparing a DC motor and three-phase asynchronous motor (Fig. 2.35), highlights the problems. In the DC, the
42、values that are important for generating torque flux () and armature current are fixed with respect to size and phase position, based on the orientation of the field windings and the position of the carbon brushes (Fig. 2.35a). In a DC motor the armature current and flux-generating cur- rent are at
43、right angles and neither value is very high. In an asynchronous motor the position of the flux () and the rotor current I1depends on the load. Furthermore unlike a DC motor, the phase angles and current are not directly measurable from the size of the stator. Using a mathematical motor model, the to
44、rque can, however, be calculated from the relationship between the flux and the stator current. U ILIM IM IS I M I sinG GD I I a)b) Ui Fig. 2.35Comparison between DC and AC asynchronous machines DC machine Simplified vector diagram of asyn- chronous machine for one load point 92CHAPTER2: FREQUENCY C
45、ONVERTERS The measured stator current (IS) is separated into the compo- nent that generates the torque (IL) with the flux ()at right angles to these two variables (IB). These generate the motor flux (Fig. 2.36). Using the two current components, torque and flux can be influ- enced independently. How
46、ever, as the calculations, which use a dynamic motor model, are quite complicated, they are only financially viable in digital drives. As this technique divides the control of the load-independent state of excitation and the torque it is possible to control an asynchronous motor just as dynamically
47、as a DC motor pro- vided you have a feedback signal. This method of three-phase AC control also offers the following advantages: good reaction to load changes precise speed regulation full torque at zero speed performance comparable to DC drives. L T IS L sin IM U IW IB IS Fig. 2.36Calculation of th
48、e current components for field-oriented regulation :Angular velocity IS: Stator current IB: Flux-generating current IW: Active current/rotor current L: Rotor flux V/f characteristic and flux vector control The speed control of three-phase AC motors has developed in recent years on the basis of two d
49、ifferent control principles: normal V/f or SCALAR control, and flux vector control. Both methods have advantages, depending on the specific requirements for drive performance (dynamics) and accuracy. V/f characteristic control has a limited speed regulation range of approximately 1:20 and at low speed, an alternative control strategy (compensation) is required. Using this technique it is relatively simple to adapt the frequency converter to the motor and the technique is robust against instantaneous loads throughout the speed range