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1、翻译部分英文原文Design Considerations for a Soft Switched, Modular 2.4MVA Medium Voltage DriveAshish Bendre, Ian Wallace, Glen LuckjiffSteve Norris, Randy Gascoigne, William E.Brumsickle, Deepak DivanSoftSwitching Technologies Corp.8155 Forsythia St. Middleton, WI 53562Email: Robert Cuzner, Wayne SchulzEato
2、n Corporation-Navy Controls3060 W. Hope Ave.Milwaukee, WI 53216A new six-phase, 2.4 MVA, soft-switched, medium voltage drive system utilizing series stacked modules with low voltage devices has been developed. The drive system combines a new soft-switched DC-DC converter with resonant DC link invert
3、er technology to deliver extremely low THD sinusoidal output, high power density and high efficiency. The series stacked configuration with the associated single-phase loading lead to unique power and control design challenges. Device selection, control of parasitic elements, sensing methods for con
4、verter control, custom magnetic component design and clamping techniques have lead to a substantial improvement in device voltage utilization. The dc-dc converter controls must regulate the intermediate dc bus voltage under single phase loading while balancing transformer excitation and maintaining
5、zero voltage switching, among other tasks. Proper control of the RDCL inverter requires the selection and tuning of the appropriate modulator and understanding its affect on the power circuit ratings. I. INTRODUCTIONA new six phase, 2.4 MVA, soft-switched, medium-voltage drive system that features e
6、xtremely low THD, high power density and high efficiency has been developed in a collaborative effort. The drive is powered from a single 700-900Vdc source while the phase output is 1380V line-neutral and 286A rms. The application requires high power density and high efficiency, so the power convers
7、ion be done at high frequencies. High voltage (2400V) devices, which have significantly high switching losses, could not be used in the design, as they would violate the efficiency and power density targets. Instead, three inverters using commonly available lower voltage devices at 1200V, producing
8、460V rms at 286A rms were connected as a series stack to produce the required output voltage. As the input is a single uncontrolled source, a new loss-limited dc-dc converter module was developed to provide isolated, regulated dc voltage to the inverter modules.For the output stage, hard-switched PW
9、M inverters with interleaved switching have been shown to achieve low THD 1. However, the synchronization of control coupled with the higher switching losses makes this approach unattractive. Three-phase resonant dc link (RDCL) inverter modules that provide high efficiency and power density along wi
10、th a spread spectrum noiseband, which allows independent (asynchronous) operation with extremely low THD have been previously developed 2. For this work, these modules were converted to single phase and substantially modified to further improve power density and device utilization. Each output phase
11、 of the drive system contains three series connected, single-phase RDCL inverters each powered by an isolated dc- dc converter as shown in Figure 1.The high power, high frequency series stacked configuration and the application requirements lead to unique design challenges and tradeoffs for both the
12、 inverter and the DC-DC converter modules. This topology leads to single phase loading on the output of the DC-DC converter, utilizes the devices closer to their ratings and increases voltage stress on isolation boundaries. These issues affect device selection, magnetic component design, control of
13、parasitic elements, capacitor and sensor selection for both soft-switched converters; these are some of the major design issues discussed in this paper. The function of the DC-DC converter is to regulate the output dc bus voltage, while handling single-phase current loading, balancing transformer ex
14、citation, and maintaining zero voltage switching. This is accomplished by varying the operating frequency from 20-30kHz using state machine control. Control of the RDCL inverter involved a tradeoff between designing the modulator to produce low THD waveforms and rating the power circuit component to
15、 achieve high power density. II. DC-DC CONVERTER DESIGNThe major restrictions to higher frequency, high power DC-DC converters are power device switching loss, throughput loss due to transformer leakage and diode reverse recovery loss. The new DC-DC converter module uses low leakage coaxial wound tr
16、ansformers along with a novel primary commutation scheme that limits the switching loss, a split secondary that permits the use of 1200V devices and an energy recovery clamp for diode recovery 3. The converter topology is shown in Figure 2.Operation: Output voltage and current are controlled by phas
17、e shifting one leg Q1, Q2 of the primary H-bridge with respect to the other leg Q3, Q4. The turn-off switching loss in the lagging leg devices Q1, Q2 is reduced by adding resonant capacitors across them to limit the dv/dt. The turn-on loss is negligible for Q1, Q2 as the devices are turned on with t
18、heir anti-parallel diodes conducting. The turn-off loss of the leading leg devices Q3, Q4 is limited to the energy stored in the (small) leakage inductance of the transformers. The turn on loss of Q3, Q4 is lowered due to the snubbing action of the leakage inductance. Coaxial transformers with low l
19、eakage mitigate the lost duty cycle issue. These are connected in a parallel-primary, series-secondary arrangement that ensures sharing of both voltage and current. Two full wave rectifier bridges D1-D4 and D5-D8 rectify the secondary voltage and inductors Lf1, Lf2 and capacitors Cf1, Cf2 serve as t
20、he output filter. The clamp arrangement consists of clamp diodes D9, D10 and clamp capacitor Ccl. The reverse recovery energy of bridges D1-D4 and D5-D6 is transferred to clamp capacitor Ccl and hence the output by diodes D10 and D9 respectively. Device selection: Due to the high input voltage (900V
21、), 1400V IGBTs were considered. When these devices were characterized under zero voltage switching conditions 4, the turn-off loss was seen to be significantly higher than 1200V devices. Table 1 lists the measured turn-off loss in mJ for 1200V and 1400V punch through devices with a 400A current rati
22、ng. Switching waveforms are shown at a sample point for each device in Figure 3.To preserve efficiency, 1200V devices were used with a staged clamping scheme that kept the device voltage overshoots under tight control. The split output bridges and energy recovery clamp allow operation with 1200V fas
23、t recovery diodes, which have significantly better recovery characteristics compared to 1700V devices required by a single output bridge rectifier.Coaxial transformer design: Conventional transformers have high leakage inductance as compared to coaxially wound transformers. At high currents and freq
24、uencies, the time required to reverse the current in conventional transformers is longer. This reduces the maximum duty cycle and so the turns ratio has to be adjusted to achieve the same output voltage. This leads to higher currents on the primary side and greater loss. Coaxial transformers reduce
25、this problem significantly as they have controlled, low leakage inductance 5. However, multi turn designs are very difficult to manufacture. The construction of these DC-DC transformers is simplified by using stacked primary turns instead of concentric turns. The secondary is wound inside the primar
26、y and is fully coaxial with the primary. The primary uses square copper tubing to increase the fill factor. The secondary consists of parallel litz wire to limit conduction and proximity loss due to high frequency currents. While coaxial transformers have traditionally been built using cut toroidal
27、cores, it is difficult to reliably clamp these cores together, base plate mounting is mechanically challenging and thermal transfer is poor. This transformer design uses E-E cores that do not need to be cut and can be placed directly on the base plate, to aid in cooling. The use of two transformers
28、further improves the heat transfer by increasing the available surface area.III. DC-DC CONVERTER CONTROL.A new peak current controller that maintains zero voltage switching and an outer loop controller that regulates the output voltage under single phase loading are the salient aspects of the DC-DC
29、converter controls. A. Inner Loop Controller: The inner loop controller provides peak current control, zero voltage turn-on of master leg devices, frequency variation to maintain synchronous operation and cycle-by-cycle volt-second balancing when current mode control is lost. A simplified diagram of
30、 the DC-DC converter is shown in Figure 4.Traditional phase shift H-bridge DC-DC converter devices operate at a fixed frequency and at a 50% duty cycle. The applied volt-seconds are controlled by phase shifting the switching of one leg in the H-bridge with respect to the other. The normal switching
31、cycle would go as follows: Q1 ON, Q3 ON: Active state = Q2 ON, Q3 ON: Zero state = Q2 ON, Q4 ON: Active state = Q1 ON, Q4 ON: Zero state.In the new loss limited DC-DC converter 3, the losses in the slave leg devices (Q3, Q4) are limited by controlling transformer leakage inductance. To limit turn-of
32、f switching loss in the master leg, resonant capacitors are added to limit the dv/dt across the devices in the master leg (Q1-Q2). The new controller realizes loss less turn on of the master leg device Q1 (Q2) by delaying it until the anti-parallel diode D1 (D2) is conducting. The addition of resona
33、nt capacitors introduces a load dependency in master device commutation as the turn-off dv/dt is proportional to the load current; hence the delay introduced by the controller between turn off of Q1 (Q2) and turn on of Q2 (Q1) varies with load current as shown in Figure 5. For fixed frequency operat
34、ion at light load and high duty cycles, this delay can be very long resulting in the slave device Q4 (Q3) being turned on before Q2 (Q1). This leads to commutation failure on the master devices and the converter slips into a half-bridge mode with Q1 and Q2 always off. The new peak controller delays
35、the slave leg switching until the master leg switching is complete, maintaining synchronous operation by reducing the effective frequency. To reduce the transformer core size, a minimum limit on this frequency is required which is achieved by limiting the time interval between turn on of Q3 (Q4), an
36、d turn off of Q1 (Q2). Any offset in the magnetizing current creates a corresponding offset in the volt seconds applied to the transformer of opposite polarity and helps regulate the flux in the transformer. The control scheme is implemented as a high-speed state machine in a Field Programmable Gate
37、 Array (FPGA).B. Output DC bus control: The device gate signals are generated using the inner peak current controller discussed previously. Outer voltage and current loop generate the peak current command for the inner loop as shown in Figure 6. The DC-DC converter sees single-phase loading, which t
38、ranslates into 2 per unit (p.u.) load current. Due to the power density constraints, the converter devices were sized to handle 1 p.u. load. For an output load range of 0.0-0.5 p.u., the converter operated in voltage regulation mode. Beyond this power point, the converter alternated between output v
39、oltage and current mode every half cycle of the single-phase load current. The transition between the two modes contained significant hysteresis, which compensated for the loss of output voltage regulation every half cycle. The load current can be used as a feed forward term into the peak current co
40、ntroller to improve dynamics. The outer control loops are coded into a Digital Signal Processor (DSP).中文译文模块化2.4MVA中电压驱动软交换设计的注意事项阿希什Bendre，伊恩华莱士，格伦Luckjiff史蒂夫诺里斯，兰迪加斯科因，威廉E. Brumsickle，迪帕克迪旺软交换技术股份有限公司8155连翘圣米德尔顿，WI53562一个新的六相2.4兆伏安软开关的中压利用串联堆叠模块具有低电压驱动系统装置已经研制成功。该驱动系统结合了一个新的谐振直流环节逆变软开关DC-DC转换器技术，高
41、功率密度和高效率的提供极低的THD正弦波输出。该系列产品堆积与相关联的单相负载导致的配置独特的权力和控制的设计挑战。设备选型、控制的寄生元件、传感方法转换器控制、定制磁性元件设计和夹紧技术导致大幅改善设备电压利用率。中间直流总线电压下的单相负载平衡变压器的激发和保持零电压开关等任务在DC-DC转换器控制过程中必须规范。适当控制RDCL变频器需要适当的选择和调整调制器并了解其电源电路的影响率。引言一个新的六相2.4兆伏安软开关在中压具有极低的总谐波失真（THD）和高电压驱动系统已经制定了一个功率密度和高效率合作方式。用相位输出1380V线电压和286A有效值的单一的700 - 900VDC源启动
42、该驱动器。该驱动器需要高功率密度和高效率合作使电源做高频率转换。高电压（2400V）的设备有显着高的开关损耗不能被用在设计，因为他们违反了效率和功率密度目标。应该使用常用的电压在1200V的三相逆变器设备，生产286A460V RMS有效值作为一系列堆栈连接以产生所需的输出电压。当输入是单一的控制源，开发新的损失限定的dc-dc转换模块是为了隔离稳压直流电压的逆变器模块。在输出阶段，硬开关PWM逆变器用交错的开关实现低THD1。然而，同步控制再加上较高的开关损耗这种方法毫无吸引力。三相谐振直流环节逆变器（RDCL）模块提供高效率和功率密度随的扩频频谱噪声频带，它允许异步与极低THD 2。对于这
43、项工作，这些模块转换为单相，并大幅修改进一步提高功率密度和设备利用率。每的驱动系统的输出相包含三个系列连接，RDCL单相逆变器由一个隔离dc-dc转换器，如在图1中示出。高功率和高频率系列堆叠配置和应用的要求导致逆变器和DC-DC的挑战和权衡转换器模块设计独特。这种拓扑结构导致单相装上的DC-DC转换器的输出，利用接近他们的额定值和增益电压应力的设备隔离边界。这些问题影响设备的选择、磁性元件的设计、控制寄生元件、两个软交换的电容和传感器的选择转换器，这些都是本文所讨论的一些设计的主要问题。DC-DC转换器的功能是在调节输出直流总线电压的同时处理单相电流负载、平衡变压器励磁并维持零电压开关。这是
44、通过20-30kHz的不同工作频率的使用状态机器控制。 RDCL逆变器的控制涉及之间的折衷设计的调制器，以产生低总谐波失真波形和打分电源电路元件实现高功率密度。DC-DC转换器的设计主要限制高频率、高功率DC-DC转换器的功率器件的开关损耗，由于变压器漏感和二极管反向通量损失恢复损耗。新的DC-DC转换器模块利用一种新型低泄漏同轴电缆缠绕变压器主转相方案限制开关损耗，分裂二次，允许使用1200V的设备和能量回收钳位二极管恢复3。该转换器拓扑结构如图2所示。操作方法：输出电压和电流控制的移相一条支路Q1、Q2的主H桥相对于另一条支路Q3、Q4。导通断开关损耗的滞后桥臂器件Q1，Q2减少了它们之间
45、增加谐振电容器的dv / dt限制。设备开启反并联二极管导通，导通损耗是微不足道的Q1和Q2。领先的支路元件Q3和Q4的关断损耗是有限的能量存储在变压器的漏电感（小）。开通Q3和Q4的亏损降低是由于滞后的漏感。同轴变压器具有低泄漏减轻损失的占空比问题。这些系列号上安排的连接主要是并联以确保电压和电流两者的共享。两个全波整流桥D1-D4和D5-D8矫正二次电压和电感LF1、LF2而电容CF1、CF2作为输出滤波器。钳位电路由钳位二极管D9、D10和钳位电容CCL。桥接D1-D4和D5-D6的反向恢复的能量被转移到钳位电容器CCL，因此分别由二极管D10和D9的输出。设备选型：输入电压（900V）
46、，1400V的IGBT进行审议。在零电压开关条件下4，这些设备显着高于1200V器件的关断损耗。表1列出了测量的关断损耗在MJ1200V和1400V的额定电压与400A额定电流的设备。如图3中的每个设备在采样点处的开关波形。为了保持效率，在严格控制下使用分阶段钳位电路保持了1200V的设备的超调电压。，相比1700V器件，分开输出1200V的桥梁和能量回收钳位二极管由单一的输出整流桥运行有更好的恢复特性。同轴变压器的设计：与传统变压器相比，同轴缠绕变压器漏感高。高电流和频率的反向电流在常规变压器所需的时间较长。这降低了最大占空比和匝数比的调整以达到相同的输出电压。这将导致更高的电流在原边损失更
47、大。同轴变压器显着减轻这个问题，因为他们可控性和低漏感，5。然而，多圈缠绕的变压器非常难以制造。这些DC-DC变压器可以通过采用堆叠初级匝数来简化，而不是同心圈数。该次级线圈的原边和副边内部缠绕完全同轴。主要使用方铜管充当线轴。由于高频电流传导和接近的损耗采用次级线圈为平行绞合线。虽然同轴变压器传统上一直采用切割的环形磁芯，但它的导热能力差而且难以可靠地夹紧轴心基板这对安装具有挑战性。此变压器的设计使用EE磁芯不需要被削减，可以直接放置在底板上以帮助冷却。两个变压器的使用通过增加有效表面积进一步提高了热传递。DC-DC转换器的控制新的峰值电流控制器有零电压开关和外回路控制器，它可以很好的进行单
48、相负载下的输出电压的DC-DC转换器控制。A内环路控制器: 内环路控制器可提供峰值电流控制是零电压即可导通的主器件。它可以保持频率的变化同步动作。当电流模式控制丢失时逐周期的保持伏- 秒平衡。图4中所示的DC-DC转换器的简化框图。传统的相移的H-桥DC-DC转换器的器件以50的占空比工作在一个固定的频率。控制的移相的切换的一腿的H桥的其他方面的应用的伏-秒。该正常开关周期如下：Q1、Q3导通：活跃状态= Q2、Q3导通：零状态= Q2、Q4导通：活跃状态= Q1、Q4导通：零状态。新的损耗限制的DC-DC转换器3 通过控制变压器的漏感在从属装置（Q3，Q4）的损失是有限的。要限制关断开关损耗在主电路，安装谐振电容以限制dv / dt的整个主站内的设备（Q1-Q2）。新的控制器实现利用了在导通的主电路器件Q1（Q2）的延迟损耗少的反并联的二极管D1（D2）进行。谐振电容的另外引入的负载依赖于主设备的换向关断的dv / dt负载电流是成比例的，因此由控制器之间Q1（Q2）的关闭和开启的Q2（Q1引入的延迟）的负载电流如在图5中示出。对于固定频率工作在轻负载和高占空比的情况，这种延迟很可能会导致从器件Q4（Q3）在Q2（Q1）前被打开。这会导致主设备和半桥Q1和Q2始终关闭模式转换器进而换相失败。